WO2009101315A1 - Power weighting of a multicarrier signal on reception in a communication system - Google Patents

Power weighting of a multicarrier signal on reception in a communication system Download PDF

Info

Publication number
WO2009101315A1
WO2009101315A1 PCT/FR2009/050155 FR2009050155W WO2009101315A1 WO 2009101315 A1 WO2009101315 A1 WO 2009101315A1 FR 2009050155 W FR2009050155 W FR 2009050155W WO 2009101315 A1 WO2009101315 A1 WO 2009101315A1
Authority
WO
WIPO (PCT)
Prior art keywords
analog signal
analog
signal
amplitude
sum
Prior art date
Application number
PCT/FR2009/050155
Other languages
French (fr)
Inventor
Gautier Avril
Fabienne Moulin
Ahmed Zeddam
Original Assignee
France Telecom
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by France Telecom filed Critical France Telecom
Publication of WO2009101315A1 publication Critical patent/WO2009101315A1/en

Links

Classifications

    • HELECTRICITY
    • H03ELECTRONIC CIRCUITRY
    • H03GCONTROL OF AMPLIFICATION
    • H03G3/00Gain control in amplifiers or frequency changers without distortion of the input signal
    • H03G3/20Automatic control
    • H03G3/30Automatic control in amplifiers having semiconductor devices
    • H03G3/3052Automatic control in amplifiers having semiconductor devices in bandpass amplifiers (H.F. or I.F.) or in frequency-changers used in a (super)heterodyne receiver
    • H03G3/3078Circuits generating control signals for digitally modulated signals
    • HELECTRICITY
    • H03ELECTRONIC CIRCUITRY
    • H03MCODING; DECODING; CODE CONVERSION IN GENERAL
    • H03M1/00Analogue/digital conversion; Digital/analogue conversion
    • H03M1/12Analogue/digital converters
    • H03M1/18Automatic control for modifying the range of signals the converter can handle, e.g. gain ranging
    • H03M1/186Automatic control for modifying the range of signals the converter can handle, e.g. gain ranging in feedforward mode, i.e. by determining the range to be selected directly from the input signal
    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04BTRANSMISSION
    • H04B3/00Line transmission systems
    • H04B3/54Systems for transmission via power distribution lines
    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04LTRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00Modulated-carrier systems
    • H04L27/26Systems using multi-frequency codes
    • H04L27/2601Multicarrier modulation systems
    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04LTRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00Modulated-carrier systems
    • H04L27/26Systems using multi-frequency codes
    • H04L27/2601Multicarrier modulation systems
    • H04L27/2614Peak power aspects
    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04LTRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00Modulated-carrier systems
    • H04L27/26Systems using multi-frequency codes
    • H04L27/2601Multicarrier modulation systems
    • H04L27/2647Arrangements specific to the receiver only
    • H04L27/2649Demodulators
    • H04L27/2653Demodulators with direct demodulation of individual subcarriers

Definitions

  • the present invention relates to a power weighting of a multicarrier signal on reception in a communication system.
  • the present invention relates to a power frequency weighting of a multicarrier signal.
  • the communication system supports a multi-carrier modulation of digital signals according to OFDM orthogonal frequency division multiplexing (Orthogonal Frequency Division Multiplexmg).
  • a known multicarrier digital communication system comprises a transmission device DT and a reception device DR.
  • An OFDM type multicarrier digital communication consists in transmitting an initial digital signal SNI including data in a wide frequency band from the transmission device DT.
  • a modulator MD in the device DT modulates the bits of the signal SNI into complex symbols, for example according to a modulation of amplitude and phase QAM.
  • the symbols are grouped in packets of I successive symbols.
  • the I symbols of each group are put in parallel to form I digital components CNI ⁇ ,
  • the MI FFT module performs a fast IFFT (Inverse Fast Fourier Transform) Fourier transform, ie a frequency-time conversion, for modulating I carriers with narrow frequency bands of the same width respectively by the I parallel symbols of each group.
  • IFFT Inverse Fast Fourier Transform
  • Each digital component is thus associated with a respective carrier, also called "sub-carrier”.
  • I 1 / I modulated carriers undergo parallel-serial conversion to the symbol frequency to be combined into a time-domain digital signal SN T. The latter comprises a digital time-level period of each corresponding symbol has a sum of symbols on the carriers.
  • the digital time signal is converted into an analog broadband multicarrier signal SA_T1 by means of a digital-to-analog converter CNA and is optionally amplified by an amplifier.
  • a coupler CP_T adapted to a transmission channel CT of a communication network connecting a plurality of communication devices between them transmits the amplified analog signal SA_T2 which is transmitted to a reception device DR of a second communication system via the transmission channel CT.
  • Such a communication network linking a transmission device of a first communication system to a receiving device of a second communication system may be a telephone network, for example of the ADSL ("Asymmetric Digital Subscriber Lme") type, a radio network for example for broadcasting type DAB (in English "Digital Audio Broadcastmg") or for broadcast type DVB (Digital Video Broadcast "in English), or a communication network on power line type CPL
  • the CT transmission channel for each of these examples is then: the copper pair for the telephone network, the air for the radio network and the power lines for the power line systems.
  • the transmitted signal SA_T2 generally undergoes disturbances.
  • the received analog signal SA_R1 by the receiving device DR is shaped in a reception coupler CP_R adapted to the channel CT and comprising a low noise amplifier for amplifying the shaped signal.
  • the signal SA_R2 at the output of the coupler CP_R is converted into a digital signal SN R in a CAN-to-digital converter.
  • the latter After having been paralleled on I input channels of a Fourier analyzer module M F p ⁇ , the latter performs a Fast Fourier Transform (FFT), that is to say a conversion time-frequency, to demodulate the I carrier I symbols parallel to each symbol period and thus recover I digital components CNF] _, CNF 1 , CNF 1 .
  • FFT Fast Fourier Transform
  • the analog-to-digital converters CAN have a limited operating voltage range, for example between -1 V and +1 V. If the amplitude of the signal SA R2 coming from the coupler CP R exceeds this voltage range and is for example included in FIG. the voltage range [-3 V, +3 V], the digital signal SN_R is then outputted, thus generating errors conversion. Conversely, if the amplitude of the SA_R2 signal from the coupler is less than the operating voltage range of the CAN converter and is for example in the voltage range [- 0.2 V, +0.2 V], a part the operating range of the CAN converter can not be used, which increases the impact of the quantization noise of the CAN converter.
  • the quantization noise corresponds to the floor noise of the CAN converter. The level of this noise depends on the number of bits of the converter and the voltage range on which it operates.
  • an Automatic Gain Control (AGC) gain controller (AGC) between the output of the receive coupler CP_R and the input of the analog-to-digital converter CAN first adjusts the amplitude of the received analog signal SA_R2 to the operating range of the CAN converter to obtain a suitable signal SA GV to the operating range of the CAN converter. This operation does not modify the signal / noise ratio received, the actual noise received being amplified or attenuated in the same way as the useful signal received over the entire frequency band.
  • the signal-to-noise ratio seen by the system on each carrier i of the received signal SA_R2 can be expressed according to the following relation, with 1 ⁇ i ⁇ I:
  • Br 1 the real noise of the signal on the carrier i, induced by the transmission channel CT, ⁇ GCAG: I e coefficient of amplification of the AGC controller, and
  • the amplification coefficient G QAG depends on the total amplitude of the signal SA_R2 received by the coupler. The higher the amplitude of this signal, the less the coefficient G ⁇ AG is.
  • Laboratory tests on PLC systems have shown that the noise of the CAN converter is mainly greater than the real noise of the CT transmission channel. Most communication systems, because of this limitation of the CAN converter, operate below the capacity of the transmission channel.
  • the amplitude of the received signal SA_R2 is largely related to the signal power on these carriers. Consequently, the AGC controller amplifies the received signal less because of the very high power level of these few carriers. This then implies a decrease in the signal-to-noise ratio including on the lower power carriers, and thus a reduction in the rate that it is possible to transmit on these carriers.
  • the present invention overcomes the drawbacks mentioned above by a method of receiving a multicarrier analog signal by a reception device comprising a converter analog / digital.
  • the method is characterized in that it comprises the steps of:
  • the amplitudes of the carriers of the received signal are thus weighted selectively to be adjusted optimally taking into account the characteristics of the analog-digital converter of the receiving device. These weights are determined to increase the total throughput by decreasing the quantization noise introduced by the analog-to-digital converter during the conversion of the summed analog signal.
  • the invention makes it possible, without increasing the complexity of the remainder of the system, to increase the real data rate of the data signal.
  • the carriers of the analog signal have frequencies selected by the N simultaneous bandpass filtering as a function respectively of N fixed frequency bands in order to produce the N filtered signals, and the N gains applied respectively to the filtered signals are variable and determined according to a maximum total bit rate of the summed analog signal and the amplitude of the summed analog signal adapted to the amplitude threshold.
  • the carriers of the analog signal have frequencies selected according to N simultaneous bandpass filtering as a function respectively of N bands of variable frequency determined according to respective fixed gains to be applied to the filter signals, the maximum total rate of the sum analog signal and the amplitude of the analog signal sum adapted to the amplitude threshold.
  • the invention also has forcget a device for receiving a multicarrier analog signal comprising an analog / digital converter.
  • the device is characterized in that it further comprises:
  • N bandpass filters for filtering the analog signal respectively in N analog filter signals each including at least one carrier, N power weighting means for weighting N respectively the N analog filter signals, and an adder for summing the weighted analog filter signals into an analog signal sum whose amplitude is adapted to an amplitude threshold.
  • the N bandpass filters are capable of filtering the carriers of the analog signal as a function respectively of N fixed frequency bands in order to produce the N filter signals, the N powerweighters respectively have N gains. variables to be applied to the filter signals.
  • the device further comprises a gain controller for determining the N variable gains as a function of a maximum total rate of the analog sum signal and the amplitude of the analog sum signal adapted to the amplitude threshold.
  • the N bandpass filters are able to filter the carriers of the analog signal by function respectively of N variable frequency bands.
  • the device further comprises a filter controller for determining the N variable frequency bands according to respective fixed gains to be applied respectively to the filter signals by the N power weighting (Pd] _- Pd ⁇ ), the maximum total throughput of the filter. sum signal and the amplitude of the analog signal sum adapted to the amplitude threshold.
  • FIG. 1 is a schematic block diagram of a multicarrier communication system as discussed above;
  • FIG. 2 is a schematic block diagram of a reception device according to a first embodiment of the invention;
  • FIG. 3 is an algorithm of a variable gain determination process in a gain controller of the reception device.
  • FIG. 4 is a schematic block diagram of a receiving device according to a second embodiment of the invention.
  • the invention is applicable to networks and PLC communication systems, acronym for line carrier current. Nevertheless, the invention can be generalized to all OFDM networks and communication systems (Orthogonal Frequency Multiplexmg division (in English) and also to networks and broadband communication systems of UWB type ("Ultra WideBand" in English).
  • a reception device DRa comprises, in a similar manner to the known reception device shown in FIG. 1, a coupler CP receiving since the CT transmission channel an analog multi-carrier signal of the OFDM type, an analog-digital converter ADC, a quick Fou ⁇ er transformation module M FT ⁇ e t a demodulator DM.
  • a power weighting device DPP is interconnected between the output of the CP coupler and the input of the CAN-to-digital converter. The device DPP processes at the output of the coupler CP the analog multicarrier signal SA which has possibly undergone in the coupler CP a low gain amplification.
  • the analog signal SA comprises I carriers Pt ⁇ a Ptj which have been subjected to modulations of different amplitudes and phases.
  • Each carrier Pt 1 with 1 ⁇ i ⁇ I, corresponds to at least a component of the digital signal associated with a carrier frequency Fp 1 .
  • the power weighting device DPP comprises a power distributor RP receiving the analog signal SA, N selection stages Ft n -Pd n each comprising a bandpass filter Ft n and a weighting device of power Pd n , with 1 ⁇ n ⁇ N, and an adder SM to obtain at the output of the DPP device an analog signal sum SAS with carriers having weighted powers.
  • the entities RP, Fti to Ft N , Pd 1 to Pd N and SM are represented in the form of functional blocks, most of which provide functions relating to the invention and may correspond to software and / or hardware modules.
  • the power distribution unit RP distributes the power of the analog signal SA which delivers by the coupler CP in N analog signals SA 1 to SA 1 respectively to the inputs of the N stages of selection.
  • Each analog signal SA n comprises the I modulated carriers Pt ⁇ -Ptj of the signal SA.
  • the bandpass filter Ft n selects in the respective analog signal SA n one or more carrier frequencies according to a selection relating to the first embodiment of the invention, or to the second embodiment of the invention described later.
  • a filter signal SAF n at the output of the filter FT n only includes the carriers relative to the frequencies selected by the filter.
  • the power weighting device Pd n applies to the filter signal SAF n a variable gain GV n attenuating or amplifying the power of the carriers in the signal SAF n in order to adjust the amplitude of the signal SAF n as a function of the amplitudes of the other filter signals SAF ⁇ .
  • Each power weighting instrument Pd n provides an SAP n weighted signal whose amplitude of the selected carriers has been adjusted.
  • the summator SM sum the weighted filter signals SAP 1 to SAP N into a single signal sum SAS comprising all the carriers Pti ⁇ Pt N.
  • the sum analog signal SAS provides a maximum total bit rate and has an amplitude smaller than the amplitude limit LIMQ ⁇ N ⁇ U CAN converter to which the SAS signal is applied to be converted into a signal numeric SN_R.
  • each bandpass filter Ft of the weighting device selects the carriers of the signal SA n having frequencies comprised in a respective fixed frequency band delimited by a low cutoff frequency Fcb n and a high cutoff frequency Fch n .
  • the frequency bands of the bandpass filters Fti to Ft are substantially disjoint and contiguous two by two and together cover the frequencies of the carriers Pt 1 to Pt 1 .
  • the DPP device operates in a frequency band between 2 MHz and 28 MHz and comprises 5 bandpass filters Fti to Ft 5 having cut-off frequencies distributed according to the following table:
  • the power distributor RP and the N bandpass filters Ft ⁇ a Ft ⁇ are replaced by a frequency divider dividing the wide signal. band received in N narrow-band signals (Fcb] _, FCh 1 ) a (Fcb N , Fch N ).
  • Each carrier Pt 1 of the signal SA n at the input of the filter Ft n is associated with a selection coefficient Qn i equal to 1 if the frequency Fp 1 of the corresponding carrier is in the respective fixed frequency band (Fcb n , Fch n ).
  • the selection coefficient associates Q n ⁇ 1 is 0. Only the carriers of the signal SA n whose frequencies are associated with equal selection coefficients a 1 are present in the filter signal SAF n provided by the filter Ft n and are processed by the power weighting instrument Pd n . The other carriers are not processed by the weighting instrument Pd n .
  • a variable gain GV n is applied to the filter signal SAF n in the power weighting instrument Pd n .
  • All the weighting modulators Pd 1 to Pd N of the N stages of selection are connected by a gain control bus to a gain controller CG which determines the variable gains GVi to GV N as a function of a maximum total flow rate D Tmax and the amplitude amp (P ⁇ ) of the analog signal sum SAS to obtain at the output of the device DPP.
  • the gain controller CG is connected to or incorporates an MR memory able to record in particular all the variable gains GV ] _ a GV N determmines.
  • a method for determining the variable gains GV 1 to GV N relative to the first embodiment of the invention and implemented in the gain controller CG comprises steps E 1 to E 12.
  • the process successively assigns each of J predetermined gain values ai a otj to the gain of each of the power weighting Pd] _ a Pd N so that all combinations of predetermined gain values are tested by the gain controller CG.
  • the method comprises N recursion loops Bx a B N included in each other and respectively relating to the variations of the power factor weight gain Pdi to Pd N.
  • values of the variable gains GV] _ a GV N are selected for which the total rate D ⁇ of the SAS signal at the output of the device DPP is maximum and for which the amplitude of the SAS signal is less than the LIMCAN amplitude limit of the CAN converter.
  • FIG. 3 only 3 recursion loops Bi, B n and B N among the N loops are represented.
  • Each recurrence loop Bi, B n , B N is repeated as long as the index Ki, K n , K N associates is not equal to J + 1.
  • the last recurrence loop B N comprises the steps E4 to E8 corresponding to the determination of the variable gains GVi to GV N according to conditions specific to the step E7.
  • CG initializes in zero in the MR memory the variable gains GVi to GV N , and the maximum total flow D Tmax of the analog signal summed SAS, used later during the execution of the process.
  • step E4 in the last recurrence loop B N the gain controller CG assigns the predetermined gain values ⁇ jQ, ... ⁇ j ⁇ ; n , ... ⁇ ⁇ respectively to the gains Gi ... G n , ... G N relative to the N stages of selection, with ⁇ ] _ ⁇ i ⁇ otj, ... ai - ⁇ Kn - ⁇ J f - has the - KN ⁇ - ⁇ J
  • These gains G to G N are stored in the memory MR.
  • step E5 the controller CG determines the total signal-to-noise ratio Rsb x of each carrier Pt 1 , by adding the signal to noise ratios Rsb ⁇ / 1 to Rsb N , x of the carrier Pt 1 determined respectively for each stage. selection Ft n -Pd n according to the gain G n assigned to step E4.
  • the total signal-to-noise ratio RSb 1 for the carrier i is determined according to the following relation:
  • N the number of selection stages of the power weighting device DPP
  • PS x the power of the signal on the carrier Pt 1
  • PB 1 the power of the noise on the carrier Pt 1
  • PE 1 the sampling noise power of the CAN converter for the carrier Pt 1 ,
  • the gain controller CG also determines the total rate D 1 of a carrier Pt 1 as a function of the total ratio RSb 1 obtained.
  • step E6 the gain controller CG determines the total power P ⁇ p of the signal SAS by summing the total power Pp t x of each carrier Pt 1 defined in the N stages of selection as a function respectively of the gains Gx to G ⁇ assigned to step E4 according to the following relation:
  • the gain controller CG applies an amp function to the total power P i previously determined in order to obtain the amplitude of the summed signal SAS as a function of the gains G x at G i attributed to the step E 4.
  • step E7 if the total flow rate D ⁇ is greater than the maximum total flow D ⁇ p max and if the amplitude amp (PT) of the signal SAS is less than the amplitude limit LIMQAN of the converter CAN, then the controller of gain awards to all winnings variables GV] _ GV N respectively the values Gx a GJ ⁇ J previously allocated in step E4 and stored in the memory MR CG controller.
  • the gain controller also assigns the maximum total bit rate D Tmax ⁇ D the total flow rate determined in step E5.
  • the variable gains GV] _ a GV n and the flow rate D ⁇ max are memorized in the memory MR by overwriting the old values.
  • the Variable gain values GV] _ a GV N are considered definitively determined for a maximum total rate and a signal amplitude less than the amplitude limit of the CAN converter, and are stored in the MR memory.
  • the power weighting device Pd n applies to the filter signal SAF n the variable gain GV n stored in order to obtain a weighted filter signal SAP n .
  • the gain weightings Pd] _ a Pd N and the gain controller GC are replaced in the selection stages Ft ⁇ -Pd ⁇ to Ft N -Pd N respectively by automatic gain controllers.
  • the automatic gain controller in each selection stage Ft n -Pd n automatically adjusts the amplitude of the respective filter signal SAF n comprising only the carriers selected by the bandpass filter FT n .
  • the N automatic gain controllers have in common an amplitude adjustment threshold which must not be exceeded by the N powers of the SAP signals at SAP N. This amplitude threshold may be equal to the amplitude limit of the CAN converter. If the amplitude threshold does not correspond to the amplitude limit LIM ⁇ AN an automatic gain controller AGC is placed between the adder SM and the ADC converter to adjust the amplitude of the summed signal according to amplitude SAS LIMC AN .
  • the gains GF] _ GFjvj of the power weighting Pd] _ to Pd ⁇ are fixed and the passbands of the bandpass filters are variable in order to select in each selection stage Ft n -Pd n of the signal carriers SA n having substantially equal powers in a predetermined tolerance interval.
  • the frequency bands of the bandpass filters F 1 to F 1 are substantially disjoint and contiguous two by two and together cover the carrier frequencies Pti to Pt 1 .
  • the low and high variable cutoff frequencies FVcbi and FVCh 1 , ... FVcb n and FVCh n ⁇ ... FVcb N and FVch ⁇ bounding a variable passband of the bandpass filter Fti ... Ft n , ... Ft ⁇ are determined by a filter controller CF connected to all bandpass filters Fti to FT N.
  • Each bandpass filter Ft n selects the carriers of the signal SA n having frequencies within its bandwidth determined by the filter controller CF.
  • Each carrier Pt 1 of the signal SA n at the input of the filter Ft n is associated with a selection coefficient Qn, i equal to 1 if the corresponding carrier frequency Ft 1 is included in the passband of the filter.
  • the associated selection coefficient Q n , i is zero. Only the filtered carriers in the SA signal n associated with a selection coefficient equal to 1 are processed in the selection stage Ft n -Pd n and are present in the filter signal SAF n provided by the filter Ft n . The other components are not processed.
  • the fixed gain GF n in the power weighting Pd n is a priori different from the fixed gains in the other power weightings and is applied to the filter signal SAF n in the power weighting Pd n .
  • the fixed gain GF n serves to weight, that is to say to amplify or attenuate the amplitude of the filter signal SAF n and thus the power of the selected carriers.
  • the gain GF n attenuates the amplitude of the signal SAF n .
  • the gain GF n amplifies the amplitude of the signal SAF n .
  • the weighted signals SAPi a SAP ⁇ have substantially equal amplitudes, but are not adjusted according to the amplitude limit LIM Q AN of the digital analog converter CAN.
  • An automatic gain controller AGC is interconnected between the output of the summator SM and the input of the digital analog converter CAN in order to adjust the power of the total signal sum SAS as a function of the amplitude limit LIM CAK [of the CAN converter.
  • the method of determining the bandpass filter cut-off frequencies is implemented in the filter controller CF and is analogous to the variable gain determination process, and successively assigns each of predetermined cutoff frequency values to the frequencies. to cut each of the bandpass filters Fti to Ft ⁇ so that all combinations of predetermined cutoff frequency values are tested by the CG gain controller.
  • the method for determining the cut-off frequencies also comprises N recursive loops included in each other and relating respectively to the variations of the cut-off frequencies of the band-pass filters Ft] _ a Ft ⁇ .
  • the flow rate and the amplitude are determined in particular according to the predefined fixed gains of the N stages of selection and the powers of the filtered carriers in each selection stage according to the variable cutoff frequencies tested.
  • the selective power weighting device DPP can also operate by combining the two previously described embodiments comprising for each selection stage a bandpass filter having a variable frequency band and a power weighting applying a variable gain.
  • the selective power weighting device DPP may be integrated in the receiving device of the communication system as present in the preceding examples, or with an input coupler be distinct from it.

Abstract

A reception device (DRa) comprising an analogue-digital converter adapts the power of a multicarrier analogue signal (SA) of OFDM type so as to obtain a summed analogue signal (SAS) whose amplitude is adapted to an amplitude threshold. N bandpass filters (Ft1-FtN) filter the analogue signal respectively into N filtered analogue signals (SAF1-SAFN) each including at least one weighters (Fp1-FpI). N power weighters (Pd1-PdN) weight by N gains respectively the N filtered analogue signals. A summator (SM) sums the weighted filtered analogue signals into the summed analogue signal. The frequency bands of the filters may be fixed and the gains of the weighters may be variable, or vice versa.

Description

Pondération en puissance d'un signal multiporteuse à la réception dans un système de communication Power weighting of a multicarrier signal at reception in a communication system
La présente invention concerne une pondération en puissance d'un signal multiporteuse a la réception dans un système de communication.The present invention relates to a power weighting of a multicarrier signal on reception in a communication system.
Plus particulièrement, la présente invention est relative a une pondération en puissance de fréquences d'un signal multiporteuse. Le système de communication supporte par exemple une modulation multiporteuse de signaux numériques selon un multiplexage par répartition en fréquences orthogonales OFDM ("Orthogonal Frequency Division Multiplexmg" en anglais).More particularly, the present invention relates to a power frequency weighting of a multicarrier signal. For example, the communication system supports a multi-carrier modulation of digital signals according to OFDM orthogonal frequency division multiplexing (Orthogonal Frequency Division Multiplexmg).
En référence a la figure 1, un système de communication numérique multiporteuse connu comprend un dispositif de transmission DT et un dispositif de réception DR. Une communication numérique multiporteuse de type OFDM consiste a transmettre un signal numérique initial SNI incluant des données dans une bande de fréquence large depuis le dispositif de transmission DT. Selon le principe de l'OFDM, un modulateur MD dans le dispositif DT module les bits du signal SNI en des symboles complexes par exemple selon une modulation d'amplitude et de phase QAM. Les symboles sont groupes en des paquets de I symboles successifs. Les I symboles de chaque groupe sont mis en parallèle pour constituer I composantes numériques CNIχ,With reference to FIG. 1, a known multicarrier digital communication system comprises a transmission device DT and a reception device DR. An OFDM type multicarrier digital communication consists in transmitting an initial digital signal SNI including data in a wide frequency band from the transmission device DT. According to the principle of the OFDM, a modulator MD in the device DT modulates the bits of the signal SNI into complex symbols, for example according to a modulation of amplitude and phase QAM. The symbols are grouped in packets of I successive symbols. The I symbols of each group are put in parallel to form I digital components CNIχ,
CNI1, CNIj distinctes en entrée d'un module synthétiseur de Fouπer MIFFT. Le module MIFFT effectue une transformation de Fourier rapide inverse IFFT ("Inverse Fast Fourier Transform" en anglais), c'est-a-dire une conversion fréquence-temps, pour moduler I porteuses a bandes de fréquence étroites de même largeur respectivement par les I symboles parallèles de chaque groupe. Chaque composante numérique est ainsi associée a une porteuse respective, également appelée "sous-porteuse". Dans le module MIFFÏ 1/ les I porteuses modulées subissent une conversion parallele-serie a la fréquence des symboles pour être combinées en un signal numérique temporel SN T. Ce dernier comporte un niveau numérique temporel a chaque période des symboles correspondant a une somme des symboles sur les porteuses. Le signal numérique temporel est converti en un signal multiporteuse large bande analogique SA_T1 au moyen d'un convertisseur numeπque- analogique CNA et est éventuellement amplifie par un amplificateur. Un coupleur CP_T adapte a un canal de transmission CT d'un reseau de communication connectant plusieurs dispositifs de communication entre eux émet le signal analogique amplifie SA_T2 qui est transmis a un dispositif de réception DR d'un deuxième système de communication via le canal de transmission CT.CNI 1 , CNIj distinct input of a synthesizer module Fouπer MI FFT . The MI FFT module performs a fast IFFT (Inverse Fast Fourier Transform) Fourier transform, ie a frequency-time conversion, for modulating I carriers with narrow frequency bands of the same width respectively by the I parallel symbols of each group. Each digital component is thus associated with a respective carrier, also called "sub-carrier". In MIFF Module I 1 / I modulated carriers undergo parallel-serial conversion to the symbol frequency to be combined into a time-domain digital signal SN T. The latter comprises a digital time-level period of each corresponding symbol has a sum of symbols on the carriers. The digital time signal is converted into an analog broadband multicarrier signal SA_T1 by means of a digital-to-analog converter CNA and is optionally amplified by an amplifier. A coupler CP_T adapted to a transmission channel CT of a communication network connecting a plurality of communication devices between them transmits the amplified analog signal SA_T2 which is transmitted to a reception device DR of a second communication system via the transmission channel CT.
Un tel reseau de communication reliant un dispositif de transmission d'un premier système de communication a un dispositif de réception d'un deuxième système de communication peut être un reseau téléphonique par exemple de type ADSL ("Asymmetric Digital Subscriber Lme" en anglais), un reseau de radiocommunications par exemple pour de la radiodiffusion de type DAB (en anglais "Digital Audio Broadcastmg" ) ou pour de la télédiffusion de type DVB (Digital Video Broadcast" en anglais), ou un reseau de communication sur ligne électrique de type CPL (Courants Porteurs en Ligne) . Le canal de transmission CT pour chacun de ces exemples est alors : la paire de cuivre pour le reseau téléphonique, l'air pour le reseau de radiocommunications et les lignes électriques pour les systèmes a courant porteurs. Lors de la transmission par le canal de transmission CT, le signal émis SA_T2 subit généralement des perturbations. Le signal analogique reçu SA_R1 par le dispositif de réception DR est mis en forme dans un coupleur de réception CP_R adapte a le canal CT et comportant un amplificateur a faible bruit pour amplifier le signal mis en forme. Le signal SA_R2 en sortie du coupleur CP_R est converti en un signal numérique SN R dans un convertisseur analogique-numérique CAN. Apres avoir ete mis en parallèle sur I voies en entrée d'un module analyseur de Fourier MFpτ, ce dernier effectue une transformation de Fourier rapide FFT ("Fast Fourier Transform" en anglais), c'est-a-dire une conversion temps-fréquence, pour démoduler les I porteuses en I symboles parallèles a chaque période de symbole et ainsi récupérer I composantes numériques CNF]_, CNF1, CNF1. Ces I composantes numériques sont mises en série en sortie du module MFFT de manière a reconstituer le signal numérique de symboles. Celui- ci est démodule dans un démodulateur DM afin de produire un signal numérique final SNF a priori identique au signal numérique initial émis SNI.Such a communication network linking a transmission device of a first communication system to a receiving device of a second communication system may be a telephone network, for example of the ADSL ("Asymmetric Digital Subscriber Lme") type, a radio network for example for broadcasting type DAB (in English "Digital Audio Broadcastmg") or for broadcast type DVB (Digital Video Broadcast "in English), or a communication network on power line type CPL The CT transmission channel for each of these examples is then: the copper pair for the telephone network, the air for the radio network and the power lines for the power line systems. During transmission via the transmission channel CT, the transmitted signal SA_T2 generally undergoes disturbances. The received analog signal SA_R1 by the receiving device DR is shaped in a reception coupler CP_R adapted to the channel CT and comprising a low noise amplifier for amplifying the shaped signal. The signal SA_R2 at the output of the coupler CP_R is converted into a digital signal SN R in a CAN-to-digital converter. After having been paralleled on I input channels of a Fourier analyzer module M F pτ, the latter performs a Fast Fourier Transform (FFT), that is to say a conversion time-frequency, to demodulate the I carrier I symbols parallel to each symbol period and thus recover I digital components CNF] _, CNF 1 , CNF 1 . These digital components are put in series at the output of the module M FFT so as to reconstitute the digital signal of symbols. This is demodulated in a DM demodulator to produce a final digital signal SNF a priori identical to the initial digital signal transmitted SNI.
Les convertisseurs analogiques-numériques CAN ont une plage de tension de fonctionnement limitée, par exemple entre -1 V et +1 V. Si l'amplitude du signal SA R2 issu du coupleur CP R excède cette plage de tension et est par exemple comprise dans la plage de tension [-3 V, +3 V] , le signal numérise SN_R est alors ecrête en sortie, engendrant ainsi des erreurs de conversion. Inversement, si l'amplitude du signal SA_R2 issu du coupleur est inférieure à la plage de tension de fonctionnement du convertisseur CAN et est par exemple comprise dans la plage de tension [- 0,2 V, +0,2 V], une partie de la plage de fonctionnement du convertisseur CAN ne peut être utilisée, ce qui augmente l'impact du bruit de quantification du convertisseur CAN. Le bruit de quantification correspond au bruit plancher du convertisseur CAN. Le niveau de ce bruit dépend du nombre de bits du convertisseur et de la plage de tension sur laquelle celui-ci fonctionne.The analog-to-digital converters CAN have a limited operating voltage range, for example between -1 V and +1 V. If the amplitude of the signal SA R2 coming from the coupler CP R exceeds this voltage range and is for example included in FIG. the voltage range [-3 V, +3 V], the digital signal SN_R is then outputted, thus generating errors conversion. Conversely, if the amplitude of the SA_R2 signal from the coupler is less than the operating voltage range of the CAN converter and is for example in the voltage range [- 0.2 V, +0.2 V], a part the operating range of the CAN converter can not be used, which increases the impact of the quantization noise of the CAN converter. The quantization noise corresponds to the floor noise of the CAN converter. The level of this noise depends on the number of bits of the converter and the voltage range on which it operates.
Pour pallier ce problème, un contrôleur automatique de gain CAG (AGC "Automatic Gain Control" en anglais) entre la sortie du coupleur de réception CP_R et l'entrée du convertisseur analogique- numérique CAN ajuste au préalable l'amplitude du signal analogique reçu SA_R2 à la plage de fonctionnement du convertisseur CAN pour obtenir un signal adapté SA GV à la plage de fonctionnement du convertisseur CAN. Cette opération ne modifie pas le rapport signal/bruit reçu, le bruit réel reçu étant amplifié ou atténué de la même façon que le signal utile reçu sur toute la bande de fréquence. Le rapport signal sur bruit vu par le système sur chaque porteuse i du signal reçu SA_R2 peut être exprimé selon la relation suivante, avec 1 < i < I :To overcome this problem, an Automatic Gain Control (AGC) gain controller (AGC) between the output of the receive coupler CP_R and the input of the analog-to-digital converter CAN first adjusts the amplitude of the received analog signal SA_R2 to the operating range of the CAN converter to obtain a suitable signal SA GV to the operating range of the CAN converter. This operation does not modify the signal / noise ratio received, the actual noise received being amplified or attenuated in the same way as the useful signal received over the entire frequency band. The signal-to-noise ratio seen by the system on each carrier i of the received signal SA_R2 can be expressed according to the following relation, with 1 <i <I:
Rsb, = Cl * GCAGRsb, = C l * G CAG
Br1 G CAG + Br C,ANBr 1 G CAG + Br C, AN
avec : - C1 : la composante utile du signal sur la porteuse i,with: - C 1 : the useful component of the signal on the carrier i,
- Br1 : le bruit réel du signal sur la porteuse i, induit par le canal de transmission CT, ~ GCAG : Ie coefficient d'amplification du contrôleur CAG, etBr 1 : the real noise of the signal on the carrier i, induced by the transmission channel CT, ~ GCAG: I e coefficient of amplification of the AGC controller, and
- BΓÇAN le bruit du convertisseur analogique- numérique . En augmentant le coefficient d'amplification GCAG du contrôleur CAG, le bruit BrçAN ^u convertisseur devient négligeable.- BΓÇAN the noise of the analog-digital converter. By increasing the amplification coefficient GCAG of the controller CAG, the noise Brc AN ^ u converter becomes negligible.
Cependant le coefficient d'amplification GQAG dépend de l'amplitude totale du signal SA_R2 reçu délivre par le coupleur. Plus l'amplitude de ce signal est élevée, moins le coefficient G^AG l'est. Des tests en laboratoire sur des systèmes CPL ont montre que le bruit du convertisseur CAN est majoritairement supérieur au bruit réel du canal de transmission CT. La plupart des systèmes de communication, en raison de cette limitation du convertisseur CAN, fonctionnent en deçà de ce que permet la capacité du canal de transmission. En outre, lorsque quelques porteuses ont un niveau de puissance beaucoup plus eleve que les autres porteuses, l'amplitude du signal reçu SA_R2 est en très grande partie liée a la puissance du signal sur ces porteuses. Par conséquent, le contrôleur CAG amplifie moins le signal reçu en raison notamment du très fort niveau de puissance de ces quelques porteuses. Ceci implique alors une diminution du rapport signal/bruit y compris sur les porteuses de plus faible puissance, et ainsi une diminution du débit qu'il est possible de transmettre sur ces porteuses.However, the amplification coefficient G QAG depends on the total amplitude of the signal SA_R2 received by the coupler. The higher the amplitude of this signal, the less the coefficient G ^ AG is. Laboratory tests on PLC systems have shown that the noise of the CAN converter is mainly greater than the real noise of the CT transmission channel. Most communication systems, because of this limitation of the CAN converter, operate below the capacity of the transmission channel. In addition, when a few carriers have a much higher power level than other carriers, the amplitude of the received signal SA_R2 is largely related to the signal power on these carriers. Consequently, the AGC controller amplifies the received signal less because of the very high power level of these few carriers. This then implies a decrease in the signal-to-noise ratio including on the lower power carriers, and thus a reduction in the rate that it is possible to transmit on these carriers.
La présente invention remédie aux inconvénients évoques ci-dessus par un procède de réception d'un signal analogique multiporteuse par un dispositif de réception comprenant un convertisseur analogique/numérique. Le procédé est caractérisé en ce qu'il comprend les étapes de :The present invention overcomes the drawbacks mentioned above by a method of receiving a multicarrier analog signal by a reception device comprising a converter analog / digital. The method is characterized in that it comprises the steps of:
N filtrages passe-bande simultanés du signal analogique respectivement en N signaux analogiques filtrés incluant chacun au moins une porteuse,N simultaneous bandpass filterings of the analog signal respectively in N filtered analog signals each including at least one carrier,
N pondérations par N gains respectivement des N signaux analogiques filtrés, et une sommation des signaux analogiques filtrés pondérés en un signal analogique sommé dont l'amplitude est adaptée à un seuil d'amplitudeN weightings by N gains respectively of the N filtered analog signals, and a summation of the weighted filtered analog signals into a summed analog signal whose amplitude is adapted to an amplitude threshold
Les amplitudes des porteuses du signal reçu sont ainsi pondérées sélectivement afin d'être ajustées de façon optimale compte tenu des caractéristiques du convertisseur analogique-numérique du dispositif de réception. Ces pondérations sont déterminées pour augmenter le débit total en diminuant le bruit de quantification introduit par le convertisseur analogique-numérique lors de la conversion du signal analogique sommé. L'invention permet, sans augmenter la complexité du reste du système, d'augmenter le débit réel du signal de données.The amplitudes of the carriers of the received signal are thus weighted selectively to be adjusted optimally taking into account the characteristics of the analog-digital converter of the receiving device. These weights are determined to increase the total throughput by decreasing the quantization noise introduced by the analog-to-digital converter during the conversion of the summed analog signal. The invention makes it possible, without increasing the complexity of the remainder of the system, to increase the real data rate of the data signal.
Selon une première réalisation de l'invention, les porteuses du signal analogique ont des fréquences sélectionnées par les N filtrages passe-bande simultanés en fonction respectivement de N bandes de fréquence fixes afin de produire les N signaux filtrés, et les N gains appliqués respectivement aux signaux filtrés sont variables et déterminés en fonction d'un débit total maximal du signal analogique sommé et de l'amplitude du signal analogique sommé adaptée au seuil d'amplitude.According to a first embodiment of the invention, the carriers of the analog signal have frequencies selected by the N simultaneous bandpass filtering as a function respectively of N fixed frequency bands in order to produce the N filtered signals, and the N gains applied respectively to the filtered signals are variable and determined according to a maximum total bit rate of the summed analog signal and the amplitude of the summed analog signal adapted to the amplitude threshold.
Selon une deuxième réalisation de l'invention, les porteuses du signal analogique ont des fréquences sélectionnées selon N filtrages passe-bande simultanés en fonction respectivement de N bandes de fréquence variables déterminées en fonction de gains respectifs fixes a appliquer sur les signaux filtres, du débit total maximal du signal analogique somme et de l'amplitude du signal analogique somme adaptée au seuil d'amplitude.According to a second embodiment of the invention, the carriers of the analog signal have frequencies selected according to N simultaneous bandpass filtering as a function respectively of N bands of variable frequency determined according to respective fixed gains to be applied to the filter signals, the maximum total rate of the sum analog signal and the amplitude of the analog signal sum adapted to the amplitude threshold.
L'invention a aussi pour ofcget un dispositif de réception d'un signal analogique multiporteuse comprenant un convertisseur analogique /numérique. Le dispositif est caractérise en ce qu'il comprend en outre :The invention also has forcget a device for receiving a multicarrier analog signal comprising an analog / digital converter. The device is characterized in that it further comprises:
N filtres passe-bande pour filtrer le signal analogique respectivement en N signaux analogiques filtres incluant chacun au moins une porteuse, N pondérateurs de puissance pour pondérer par N gains respectivement les N signaux analogiques filtres, et un sommateur pour sommer les signaux analogiques filtres pondères en un signal analogique somme dont l'amplitude est adaptée a un seuil d'amplitude.N bandpass filters for filtering the analog signal respectively in N analog filter signals each including at least one carrier, N power weighting means for weighting N respectively the N analog filter signals, and an adder for summing the weighted analog filter signals into an analog signal sum whose amplitude is adapted to an amplitude threshold.
Selon une première réalisation du dispositif pour pondérer, les N filtres passe-bande sont aptes a filtrer les porteuses du signal analogique en fonction respectivement de N bandes de fréquence fixes afin de produire les N signaux filtres, les N pondérateurs de puissance ont respectivement N gains variables a appliquer sur les signaux filtres. Le dispositif comprend en outre un contrôleur de gain pour déterminer les N gains variables en fonction d'un débit total maximal du signal analogique somme et de l'amplitude du signal analogique somme adaptée au seuil d'amplitude.According to a first embodiment of the device for weighting, the N bandpass filters are capable of filtering the carriers of the analog signal as a function respectively of N fixed frequency bands in order to produce the N filter signals, the N powerweighters respectively have N gains. variables to be applied to the filter signals. The device further comprises a gain controller for determining the N variable gains as a function of a maximum total rate of the analog sum signal and the amplitude of the analog sum signal adapted to the amplitude threshold.
Selon une deuxième réalisation du dispositif pour pondérer, les N filtres passe-bande sont aptes a filtrer les porteuses du signal analogique en fonction respectivement de N bandes de fréquence variables. Le dispositif comprend en outre un contrôleur de filtre pour déterminer les N bandes de fréquence variables en fonction de gains respectifs fixes a appliquer respectivement sur les signaux filtres par les N pondérateurs de puissance (Pd]_- Pd^) , du débit total maximal du signal somme et de l'amplitude du signal analogique somme adaptée au seuil d'amplitude.According to a second embodiment of the device for weighting, the N bandpass filters are able to filter the carriers of the analog signal by function respectively of N variable frequency bands. The device further comprises a filter controller for determining the N variable frequency bands according to respective fixed gains to be applied respectively to the filter signals by the N power weighting (Pd] _- Pd ^), the maximum total throughput of the filter. sum signal and the amplitude of the analog signal sum adapted to the amplitude threshold.
D'autres caractéristiques et avantages de la présente invention apparaîtront plus clairement a la lecture de la description suivante de plusieurs réalisations de l'invention données a titre d'exemples non limitatifs, en référence aux dessins annexes correspondants dans lesquels :Other characteristics and advantages of the present invention will appear more clearly on reading the following description of several embodiments of the invention given by way of non-limiting examples, with reference to the corresponding appended drawings in which:
- la figure 1 est un bloc-diagramme schématique d'un système de communication multiporteuse commente ci-dessus ; - la figure 2 est un bloc-diagramme schématique d'un dispositif de réception selon une première réalisation de l'invention ;FIG. 1 is a schematic block diagram of a multicarrier communication system as discussed above; FIG. 2 is a schematic block diagram of a reception device according to a first embodiment of the invention;
- la figure 3 est un algorithme d'un procède de détermination de gain variable dans un contrôleur de gain du dispositif de réception ; etFIG. 3 is an algorithm of a variable gain determination process in a gain controller of the reception device; and
- la figure 4 est un bloc-diagramme schématique d'un dispositif de réception selon une deuxième réalisation de l'invention.- Figure 4 is a schematic block diagram of a receiving device according to a second embodiment of the invention.
Selon un exemple auquel on se référera par la suite, l'invention est applicable aux reseaux et systèmes de communication CPL, acronyme de Courant Porteur de Ligne. Néanmoins, l'invention peut être généralisée a tous les reseaux et systèmes de communication de type OFDM ("Orthogonal Frequency Division Multiplexmg" en anglais) et également aux reseaux et systèmes de communication a large bande de type UWB ("Ultra WideBand" en anglais).According to an example which will be referred to hereafter, the invention is applicable to networks and PLC communication systems, acronym for line carrier current. Nevertheless, the invention can be generalized to all OFDM networks and communication systems (Orthogonal Frequency Multiplexmg division (in English) and also to networks and broadband communication systems of UWB type ("Ultra WideBand" in English).
Comme montre a la figure 2 ou 4, dans le système de communication un dispositif de réception DRa selon une première ou deuxième réalisation de l'invention comprend de manière analogue au dispositif de réception connu montre a la figure 1, un coupleur CP recevant depuis le canal de transmission CT un signal multiporteuse analogique de type OFDM, un convertisseur analogique-numérique CAN, un module de transformation de Fouπer rapide M^FT et un démodulateur DM. Selon l'invention, un dispositif de pondération de puissance DPP est interconnecte entre la sortie du coupleur CP et l'entrée du convertisseur analogique- numérique CAN. Le dispositif DPP traite en sortie du coupleur CP le signal multiporteuse analogique SA ayant subi éventuellement dans le coupleur CP une amplification a faible gain. Le signal analogique SA comporte I porteuses Pt^ a Ptj qui ont ete soumises a des modulations d'amplitudes et de phases différentes. Chaque porteuse Pt1, avec 1 < i < I, correspond au moins a une composante du signal numérique associée a une fréquence porteuse Fp1.As shown in FIG. 2 or 4, in the communication system a reception device DRa according to a first or second embodiment of the invention comprises, in a similar manner to the known reception device shown in FIG. 1, a coupler CP receiving since the CT transmission channel an analog multi-carrier signal of the OFDM type, an analog-digital converter ADC, a quick Fouπer transformation module M FT ^ e t a demodulator DM. According to the invention, a power weighting device DPP is interconnected between the output of the CP coupler and the input of the CAN-to-digital converter. The device DPP processes at the output of the coupler CP the analog multicarrier signal SA which has possibly undergone in the coupler CP a low gain amplification. The analog signal SA comprises I carriers Pt ^ a Ptj which have been subjected to modulations of different amplitudes and phases. Each carrier Pt 1 , with 1 <i <I, corresponds to at least a component of the digital signal associated with a carrier frequency Fp 1 .
Le dispositif de pondération de puissance DPP, selon les deux réalisations de l'invention, comprend un répartiteur de puissance RP recevant le signal analogique SA, N étages de sélection Ftn-Pdn comportant chacun un filtre passe-bande Ftn et un pondérateur de puissance Pdn, avec 1 < n ≤ N, et un sommateur SM pour obtenir en sortie du dispositif DPP un signal analogique somme SAS avec des porteuses ayant des puissances pondérées. Les entités RP, Fti a FtN, Pd1 a PdN et SM sont représentées sous forme de blocs fonctionnels dont la plupart assurent des fonctions ayant un lien avec l'invention et peuvent correspondre a des modules logiciels et/ou matériels.The power weighting device DPP, according to the two embodiments of the invention, comprises a power distributor RP receiving the analog signal SA, N selection stages Ft n -Pd n each comprising a bandpass filter Ft n and a weighting device of power Pd n , with 1 <n ≤ N, and an adder SM to obtain at the output of the DPP device an analog signal sum SAS with carriers having weighted powers. The entities RP, Fti to Ft N , Pd 1 to Pd N and SM are represented in the form of functional blocks, most of which provide functions relating to the invention and may correspond to software and / or hardware modules.
Le répartiteur de puissance RP repartit la puissance du signal analogique SA délivre par le coupleur CP dans N signaux analogiques SA1 a SA^ respectivement aux entrées des N étages de sélection. Chaque signal analogique SAn comporte les I porteuses modulées Ptχ-Ptj du signal SA.The power distribution unit RP distributes the power of the analog signal SA which delivers by the coupler CP in N analog signals SA 1 to SA 1 respectively to the inputs of the N stages of selection. Each analog signal SA n comprises the I modulated carriers Ptχ-Ptj of the signal SA.
Dans chaque étage de sélection Ftn-Pdn, le filtre passe-bande Ftn sélectionne dans le signal analogique respectif SAn une ou plusieurs fréquences porteuses selon une sélection relative a la première réalisation de l'invention, ou a la deuxième réalisation de l'invention décrite ultérieurement. Un signal filtre SAFn en sortie du filtre FTn comprend uniquement les porteuses relatives aux fréquences sélectionnées par le filtre. Le pondérateur de puissance Pdn applique au signal filtre SAFn un gain variable GVn atténuant ou amplifiant la puissance des porteuses dans le signal SAFn afin d'ajuster l'amplitude du signal SAFn en fonction des amplitudes des autres signaux filtres SAFχ-SAFN et en fonction d'une limite d'amplitude LIMCAN du convertisseur CAN. Pour augmenter le débit global du canal, le gain a appliquer a un signal SAFn de puissance élevée est inférieur au gain a appliquer a un signal SAFn de puissance plus faible. Chaque pondérateur de puissance Pdn fournit un signal pondère SAPn dont l'amplitude des porteuses sélectionnées a ete ajustée. Le sommateur SM somme les signaux filtres pondères SAP1 a SAPN en un unique signal somme SAS comportant toutes les porteuses Pti~PtN. En sortie du dispositif de pondération de puissance DPP, le signal analogique somme SAS offre un débit total maximal et possède une amplitude inférieure a la limite d'amplitude LIMQ^N ^U convertisseur CAN auquel le signal SAS est applique pour être converti en un signal numérique SN_R.In each selection stage Ft n -Pd n , the bandpass filter Ft n selects in the respective analog signal SA n one or more carrier frequencies according to a selection relating to the first embodiment of the invention, or to the second embodiment of the invention described later. A filter signal SAF n at the output of the filter FT n only includes the carriers relative to the frequencies selected by the filter. The power weighting device Pd n applies to the filter signal SAF n a variable gain GV n attenuating or amplifying the power of the carriers in the signal SAF n in order to adjust the amplitude of the signal SAF n as a function of the amplitudes of the other filter signals SAFχ. SAF N and as a function of a LIM CAN amplitude limit of the CAN converter. To increase the overall throughput of the channel, the gain applied to a signal SAF n high power is less than the gain applied to a signal SAF n lower power. Each power weighting instrument Pd n provides an SAP n weighted signal whose amplitude of the selected carriers has been adjusted. The summator SM sum the weighted filter signals SAP 1 to SAP N into a single signal sum SAS comprising all the carriers Pti ~ Pt N. At the output of the power scaling device DPP, the sum analog signal SAS provides a maximum total bit rate and has an amplitude smaller than the amplitude limit LIMQ ^ N ^ U CAN converter to which the SAS signal is applied to be converted into a signal numeric SN_R.
En référence a la première réalisation illustrée la figure 2, chaque filtre passe-bande Ft du dispositif de pondération sélectionne les porteuses du signal SAn ayant des fréquences comprises dans une bande de fréquence fixe respective délimitée par une fréquence de coupure basse Fcbn et une fréquence de coupure haute Fchn. Les bandes de fréquence des filtres passe-bande Fti a Ft^ sont sensiblement disjointes et contigues deux a deux et recouvrent ensemble les fréquences des porteuses Pt]_ a Pt1. Par exemple le dispositif DPP travaille dans une bande de fréquence comprise entre 2 MHz et 28 MHz et comprend 5 filtres passe-bande Fti a Ft5 ayant des fréquences de coupure reparties selon le tableau suivant :With reference to the first embodiment illustrated in FIG. 2, each bandpass filter Ft of the weighting device selects the carriers of the signal SA n having frequencies comprised in a respective fixed frequency band delimited by a low cutoff frequency Fcb n and a high cutoff frequency Fch n . The frequency bands of the bandpass filters Fti to Ft are substantially disjoint and contiguous two by two and together cover the frequencies of the carriers Pt 1 to Pt 1 . For example, the DPP device operates in a frequency band between 2 MHz and 28 MHz and comprises 5 bandpass filters Fti to Ft 5 having cut-off frequencies distributed according to the following table:
Figure imgf000013_0001
Figure imgf000013_0001
En variante, le répartiteur de puissance RP et les N filtres passe-bande Ft^ a Ft^ sont remplaces par un diviseur de fréquence divisant le signal large bande reçu en N signaux a bandes étroites (Fcb]_, FCh1) a (FcbN, FchN) .As a variant, the power distributor RP and the N bandpass filters Ft ^ a Ft ^ are replaced by a frequency divider dividing the wide signal. band received in N narrow-band signals (Fcb] _, FCh 1 ) a (Fcb N , Fch N ).
Chaque porteuse Pt1 du signal SAn en entrée du filtre Ftn est associée a un coefficient de sélection Qn i égal a 1 si la fréquence Fp1 de la porteuse correspondante est comprise dans la bande de fréquence fixe respective (Fcbn, Fchn) . Dans le cas contraire, si Fp1 < Fcbn ou Fchn < Fp1, alors le coefficient de sélection associe Qn^1 est 0. Seules les porteuses du signal SAn dont les fréquences sont associées a des coefficients de sélection égaux a 1 sont présentes dans le signal filtre SAFn fourni par le filtre Ftn et sont traitées par le pondérateur de puissance Pdn. Les autres porteuses ne sont pas traitées par le pondérateur Pdn.Each carrier Pt 1 of the signal SA n at the input of the filter Ft n is associated with a selection coefficient Qn i equal to 1 if the frequency Fp 1 of the corresponding carrier is in the respective fixed frequency band (Fcb n , Fch n ). In the opposite case, if Fp 1 <Fcb n or Fch n <Fp 1 , then the selection coefficient associates Q n ^ 1 is 0. Only the carriers of the signal SA n whose frequencies are associated with equal selection coefficients a 1 are present in the filter signal SAF n provided by the filter Ft n and are processed by the power weighting instrument Pd n . The other carriers are not processed by the weighting instrument Pd n .
Un gain variable GVn est applique au signal filtre SAFn dans le pondérateur de puissance Pdn. Tous les pondérateurs de puissances Pd1 a PdN des N étages de sélection sont relies par un bus de contrôle de gain a un contrôleur de gain CG qui détermine les gains variables GVi a GVN en fonction d'un débit total maximal DTmax et de l'amplitude amp(Pτ) du signal analogique somme SAS a obtenir en sortie du dispositif DPP. Le contrôleur de gain CG est relie ou incorpore une mémoire MR propre a enregistrer notamment tous les gains variables GV]_ a GVN détermines.A variable gain GV n is applied to the filter signal SAF n in the power weighting instrument Pd n . All the weighting modulators Pd 1 to Pd N of the N stages of selection are connected by a gain control bus to a gain controller CG which determines the variable gains GVi to GV N as a function of a maximum total flow rate D Tmax and the amplitude amp (P τ ) of the analog signal sum SAS to obtain at the output of the device DPP. The gain controller CG is connected to or incorporates an MR memory able to record in particular all the variable gains GV ] _ a GV N determmines.
En référence a la figure 3, un procède de détermination des gains variables GV]_ a GVN relatif a la première réalisation de l'invention et implemente dans le contrôleur de gain CG comprend des étapes El a E12. Le procède attribue successivement chacune de J valeurs de gain prédéterminées ai a otj au gain de chacun des pondérateurs de puissance Pd]_ a PdN afin que toutes les combinaisons de valeurs de gain prédéterminées soient testées par le contrôleur de gain CG. Le procède comprend N boucles de récurrence Bx a BN incluses les unes dans les autres et relatives respectivement aux variations des gains des pondérateurs de puissance Pdi a PdN. Dans la dernière boucle exécutée BN incluse dans toutes les autres boucles, des valeurs des gains variables GV]_ a GVN sont sélectionnées pour lesquelles le débit total Dτ du signal SAS en sortie du dispositif DPP est maximal et pour lesquelles l'amplitude du signal SAS est inférieure a la limite d'amplitude LIMÇAN du convertisseur CAN. Pour simplifier la représentation de l'algorithme du procède, a la figure 3, seules 3 boucles de récurrence Bi, Bn et BN parmi les N boucles sont représentées. Une première boucle de récurrence Bl incluant toutes les autres boucles correspond aux étapes El a ElI et teste les J valeurs possibles ai a αj pour un gain Gi = CCKI du premier pondérateur de puissance Pdi avec l'indice Ki initialement a la valeur 1 et incremente a chaque récurrence de la boucle Bi d'une unité jusqu'au nombre J. Les deux autres boucles de récurrence relatives a une boucle de récurrence intermédiaire Bn aux étapes E2 a ElO et a une dernière boucle de récurrence BN aux étapes E3 a E9 testent respectivement pour le gain Gn = a^n du pondérateur de puissance Pdn avec 1 < Kn < J et pour le gain ^N = α KN du pondérateur de puissance PdN avec 1 < KN < J, les J valeurs de gain prédéterminées. Chaque boucle de récurrence Bi, Bn, BN est réitérée tant que l'indice Ki, Kn, KN associe n'est pas égal a J + 1. La dernière boucle de récurrence BN comprend les étapes E4 à E8 correspondant à la détermination des gains variables GVi à GVN selon des conditions spécifiques à l'étape E7.With reference to FIG. 3, a method for determining the variable gains GV 1 to GV N relative to the first embodiment of the invention and implemented in the gain controller CG comprises steps E 1 to E 12. The process successively assigns each of J predetermined gain values ai a otj to the gain of each of the power weighting Pd] _ a Pd N so that all combinations of predetermined gain values are tested by the gain controller CG. The method comprises N recursion loops Bx a B N included in each other and respectively relating to the variations of the power factor weight gain Pdi to Pd N. In the last loop executed B N included in all the other loops, values of the variable gains GV] _ a GV N are selected for which the total rate D τ of the SAS signal at the output of the device DPP is maximum and for which the amplitude of the SAS signal is less than the LIMCAN amplitude limit of the CAN converter. To simplify the representation of the algorithm of the proceeds, in FIG. 3, only 3 recursion loops Bi, B n and B N among the N loops are represented. A first recursion loop B1 including all the other loops corresponds to the steps El a ElI and tests the J possible values ai a αj for a gain Gi = CCKI of the first power weighting Pdi with the index Ki initially at the value 1 and incremente at each recurrence of the loop Bi from one unit to the number J. The two other recursion loops relating to an intermediate recurrence loop B n to the steps E2 to ElO and to a last recurrence loop B N in the steps E3a E9 test respectively for the gain G n = a ^ n of the power weighting Pd n with 1 <K n <J and for the gain ^ N = α KN of the power weighting Pd N with 1 <K N <J, the J predetermined gain values. Each recurrence loop Bi, B n , B N is repeated as long as the index Ki, K n , K N associates is not equal to J + 1. The last recurrence loop B N comprises the steps E4 to E8 corresponding to the determination of the variable gains GVi to GV N according to conditions specific to the step E7.
A une étape initiale EO, le contrôleur de gainAt an initial step EO, the gain controller
CG initialise à zéro dans la mémoire MR les gains variables GVi à GVN, et le débit total maximal DTmax du signal analogique sommé SAS, utilisés ultérieurement lors de l'exécution du procédé.CG initializes in zero in the MR memory the variable gains GVi to GV N , and the maximum total flow D Tmax of the analog signal summed SAS, used later during the execution of the process.
A l'étape E4 dans la dernière boucle de récurrence BN, le contrôleur de gain CG attribue les valeurs de gain prédéterminées αjQ,... αj<;n , ... α^^ respectivement aux gains Gi ... Gn,... GN relatifs aux N étages de sélection, avec α]_ < ακi ≤ otj, ... ai - αKn - αJ f- al - αKN - αJ- Ces gains Gi à GN sont mémorisés dans la mémoire MR.In step E4 in the last recurrence loop B N , the gain controller CG assigns the predetermined gain values αjQ, ... αj <; n , ... α ^^ respectively to the gains Gi ... G n , ... G N relative to the N stages of selection, with α] _ <ακi ≤ otj, ... ai - α Kn - α J f - has the - KN α - α J These gains G to G N are stored in the memory MR.
A l'étape E5, le contrôleur CG détermine le rapport signal sur bruit total Rsbx de chaque porteuse Pt1, par addition des rapports signal sur bruit Rsb±/1 à RsbN, x de la porteuse Pt1 déterminés respectivement pour chaque étage de sélection Ftn-Pdn en fonction du gain Gn attribué à l'étape E4.In step E5, the controller CG determines the total signal-to-noise ratio Rsb x of each carrier Pt 1 , by adding the signal to noise ratios Rsb ± / 1 to Rsb N , x of the carrier Pt 1 determined respectively for each stage. selection Ft n -Pd n according to the gain G n assigned to step E4.
Le rapport signal sur bruit total RSb1 pour la porteuse i est déterminé selon la relation suivante :The total signal-to-noise ratio RSb 1 for the carrier i is determined according to the following relation:
N
Figure imgf000016_0001
n-1 avec
NOT
Figure imgf000016_0001
n-1 with
PS * G * Ω - RSbn,, ≈ = r- ^PS * G * Ω - RSb n ,, ≈ = r - ^
PBn Gr Qn, i + PEPB n G r Qn, i + PE
- N : le nombre d'étages de sélection du dispositif de pondération de puissance DPP,N: the number of selection stages of the power weighting device DPP,
- PSx : la puissance du signal sur la porteuse Pt1, - PB1 : la puissance du bruit sur la porteuse Pt1,PS x : the power of the signal on the carrier Pt 1 , PB 1 : the power of the noise on the carrier Pt 1 ,
- PE1 : la puissance du bruit d'échantillonnage du convertisseur CAN pour la porteuse Pt1,PE 1 : the sampling noise power of the CAN converter for the carrier Pt 1 ,
- Gn : le gain pour l'étage de sélection Ftn-Pdn, et - Qnrχ le coefficient de sélection du filtre passe- bande Ftn pour la porteuse Pt1.- G n : the gain for the selection stage Ft n -Pd n , and - Q nr χ the selection coefficient of the bandpass filter Ft n for the carrier Pt 1 .
Le contrôleur de gain CG détermine également le débit total D1 d'une porteuse Pt1 en fonction du rapport total RSb1 obtenu.The gain controller CG also determines the total rate D 1 of a carrier Pt 1 as a function of the total ratio RSb 1 obtained.
Le contrôleur de gain CG somme tous les débits déterminés 0χ à Dj des I porteuses Pt]_ à Ptj pour déterminer le débit total O^ du signal, soit : i Dτ = ∑DX . i = lThe gain controller CG sum all the determined rates 0χ to Dj of the carriers Pt] _ to Ptj to determine the total bit rate O ^ of the signal, ie: i D τ = ΣD X. i = l
A l'étape E6, le contrôleur de gain CG détermine la puissance totale P<p du signal SAS en sommant la puissance totale Pptx de chaque porteuse Pt1 définie dans les N étages de sélection en fonction respectivement des gains Gx à G^ attribués à l'étape E4 selon la relation suivante :In step E6, the gain controller CG determines the total power P <p of the signal SAS by summing the total power Pp t x of each carrier Pt 1 defined in the N stages of selection as a function respectively of the gains Gx to G ^ assigned to step E4 according to the following relation:
I I NI I N
Pτ = ∑ PPtl = Σ ∑ PS, * Gn * Qn/ 1 . i - l i = l n = lP τ = Σ P Ptl = Σ Σ PS, * G n * Q n / 1 . i - li = ln = l
Le contrôleur de gain CG applique une fonction amp sur la puissance totale P^ précédemment déterminée afin d'obtenir l'amplitude du signal sommé SAS en fonction des gains Gx à G^ attribués à l'étape E4.The gain controller CG applies an amp function to the total power P i previously determined in order to obtain the amplitude of the summed signal SAS as a function of the gains G x at G i attributed to the step E 4.
A l'étape E7, si le débit total Dτ est supérieur au débit total maximal D<pmax et si l'amplitude amp (PT) du signal SAS est inférieure à la limite d'amplitude LIMQAN du convertisseur CAN, alors le contrôleur de gain attribue à tous les gains variables GV]_ a GVN respectivement les valeurs Gx a GJ^J attribuées précédemment a l'étape E4 et mémorisées dans la mémoire MR du contrôleur CG. Le contrôleur de gain attribue également au débit total maximal DTmax le débit total Dτ détermine a l'étape E5. Les gains variables GV]_ a GVn et le débit Dχmax sont mémorises dans la mémoire MR par écrasement des anciennes valeurs .In step E7, if the total flow rate D τ is greater than the maximum total flow D <p max and if the amplitude amp (PT) of the signal SAS is less than the amplitude limit LIMQAN of the converter CAN, then the controller of gain awards to all winnings variables GV] _ GV N respectively the values Gx a GJ ^ J previously allocated in step E4 and stored in the memory MR CG controller. The gain controller also assigns the maximum total bit rate D Tmax τ D the total flow rate determined in step E5. The variable gains GV] _ a GV n and the flow rate Dχ max are memorized in the memory MR by overwriting the old values.
Des que toutes les combinaisons des valeurs de gain prédéterminées dans les N boucles de récurrence ont ete testées, c ' est-a-dire si K^ = J+l , Kn = J+l et KN = J+l, les valeurs de gain variable GV]_ a GVN sont considérées comme définitivement déterminées pour un débit total maximal et une amplitude du signal inférieure a la limite d'amplitude du convertisseur CAN, et sont mémorisées dans la mémoire MR.As soon as all the combinations of the predetermined gain values in the N recursion loops have been tested, that is, if K i = J + 1, K n = J + 1 and K N = J + 1, the Variable gain values GV] _ a GV N are considered definitively determined for a maximum total rate and a signal amplitude less than the amplitude limit of the CAN converter, and are stored in the MR memory.
A chaque étage de sélection Ftn-Pdn, le pondérateur de puissance Pdn applique au signal filtre SAFn le gain variable GVn mémorise afin d'obtenir un signal filtre pondère SAPn.At each selection stage Ft n -Pd n , the power weighting device Pd n applies to the filter signal SAF n the variable gain GV n stored in order to obtain a weighted filter signal SAP n .
En variante, les pondérateurs de gain Pd]_ a PdN et le contrôleur de gain GC sont remplaces dans les étages de sélection Ftχ-Pdχ a FtN-PdN respectivement par des contrôleurs automatiques de gain. Le contrôleur automatique de gain dans chaque étage de sélection Ftn-Pdn ajuste automatiquement l'amplitude du signal filtre respectif SAFn comportant exclusivement les porteuses sélectionnées par le filtre passe-bande FTn. Les N contrôleurs automatiques de gain ont en commun un seuil d'ajustement d'amplitude qui ne doit pas être dépasse par les N puissances des signaux SAP]_ a SAPN. Ce seuil d'amplitude peut être égal à la limite d'amplitude du convertisseur CAN. Si le seuil d'amplitude ne correspond pas à la limite d'amplitude LIMÇAN' un contrôleur automatique de gain CAG est placé entre le sommateur SM et le convertisseur CAN afin d'ajuster l'amplitude du signal sommé SAS selon l'amplitude LIMçAN.As a variant, the gain weightings Pd] _ a Pd N and the gain controller GC are replaced in the selection stages Ftχ-Pdχ to Ft N -Pd N respectively by automatic gain controllers. The automatic gain controller in each selection stage Ft n -Pd n automatically adjusts the amplitude of the respective filter signal SAF n comprising only the carriers selected by the bandpass filter FT n . The N automatic gain controllers have in common an amplitude adjustment threshold which must not be exceeded by the N powers of the SAP signals at SAP N. This amplitude threshold may be equal to the amplitude limit of the CAN converter. If the amplitude threshold does not correspond to the amplitude limit LIMÇAN an automatic gain controller AGC is placed between the adder SM and the ADC converter to adjust the amplitude of the summed signal according to amplitude SAS LIMC AN .
En référence à la deuxième réalisation de l'invention illustrée à la figure 4, les gains GF]_ à GFjvj des pondérateurs de puissance Pd]_ à Pd^ sont fixes et les bandes passantes des filtres passe-bande sont variables afin de sélectionner dans chaque étage de sélection Ftn-Pdn des porteuses du signal SAn ayant des puissances sensiblement égales dans un intervalle de tolérance prédéterminé. Les bandes de fréquence des filtres passe-bande Ft]_ à Ft^ sont sensiblement disjointes et contigϋes deux à deux et recouvrent ensemble les fréquences des porteuses Pti à Pt1. Les fréquences de coupure basse et haute variables FVcbi et FVCh1,... FVcbn et FVChn^... FVcbN et FVch^ bornant une bande passante variable du filtre passe-bande Fti ... Ftn,... Ft^ sont déterminées par un contrôleur de filtre CF relié à tous les filtres passe-bande Fti à FTN.With reference to the second embodiment of the invention illustrated in FIG. 4, the gains GF] _ GFjvj of the power weighting Pd] _ to Pd ^ are fixed and the passbands of the bandpass filters are variable in order to select in each selection stage Ft n -Pd n of the signal carriers SA n having substantially equal powers in a predetermined tolerance interval. The frequency bands of the bandpass filters F 1 to F 1 are substantially disjoint and contiguous two by two and together cover the carrier frequencies Pti to Pt 1 . The low and high variable cutoff frequencies FVcbi and FVCh 1 , ... FVcb n and FVCh n ^ ... FVcb N and FVch ^ bounding a variable passband of the bandpass filter Fti ... Ft n , ... Ft ^ are determined by a filter controller CF connected to all bandpass filters Fti to FT N.
Chaque filtre passe-bande Ftn sélectionne les porteuses du signal SAn ayant des fréquences comprises dans sa bande passante déterminée par le contrôleur de filtre CF. Chaque porteuse Pt1 du signal SAn en entrée du filtre Ftn est associée à un coefficient de sélection Qn, i égal à 1 si la fréquence porteuse Ft1 correspondante est comprise la bande passante du filtre. Dans le cas contraire, le coefficient de sélection associé Qn, i est zéro. Seules les porteuses filtrées dans le signal SAn associées a un coefficient de sélection égal a 1 sont traitées dans l'étage de sélection Ftn-Pdn et sont présentes dans le signal filtre SAFn fourni par le filtre Ftn. Les autres composantes ne sont pas traitées.Each bandpass filter Ft n selects the carriers of the signal SA n having frequencies within its bandwidth determined by the filter controller CF. Each carrier Pt 1 of the signal SA n at the input of the filter Ft n is associated with a selection coefficient Qn, i equal to 1 if the corresponding carrier frequency Ft 1 is included in the passband of the filter. In the opposite case, the associated selection coefficient Q n , i is zero. Only the filtered carriers in the SA signal n associated with a selection coefficient equal to 1 are processed in the selection stage Ft n -Pd n and are present in the filter signal SAF n provided by the filter Ft n . The other components are not processed.
Le gain fixe GFn dans le pondérateur de puissance Pdn est a priori différent des gains fixes dans les autres pondérateurs de puissance et est applique au signal filtre SAFn dans le pondérateur de puissance Pdn. Selon sa valeur prédéfinie, le gain fixe GFn sert a pondérer, c'est-a-dire a amplifier ou atténuer l'amplitude du signal filtre SAFn et ainsi la puissance des porteuses sélectionnées. Pour une puissance élevée, le gain GFn atténue l'amplitude du signal SAFn. Inversement, pour une puissance faible, le gain GFn amplifie l'amplitude du signal SAFn.The fixed gain GF n in the power weighting Pd n is a priori different from the fixed gains in the other power weightings and is applied to the filter signal SAF n in the power weighting Pd n . According to its predefined value, the fixed gain GF n serves to weight, that is to say to amplify or attenuate the amplitude of the filter signal SAF n and thus the power of the selected carriers. For high power, the gain GF n attenuates the amplitude of the signal SAF n . Conversely, for a low power, the gain GF n amplifies the amplitude of the signal SAF n .
Les gains des étages de sélection étant fixes, les signaux pondères SAPi a SAP^ ont des amplitudes sensiblement égales, mais ne sont pas ajustes selon la limite d'amplitude LIMQAN du convertisseur analogique numérique CAN. Un contrôleur automatique de gain CAG est interconnecte entre la sortie du sommateur SM et l'entrée du convertisseur analogique numérique CAN afin d'ajuster la puissance du signal total somme SAS en fonction de la limite d'amplitude LIMCAK[ du convertisseur CAN.Since the gains of the selection stages are fixed, the weighted signals SAPi a SAP ^ have substantially equal amplitudes, but are not adjusted according to the amplitude limit LIM Q AN of the digital analog converter CAN. An automatic gain controller AGC is interconnected between the output of the summator SM and the input of the digital analog converter CAN in order to adjust the power of the total signal sum SAS as a function of the amplitude limit LIM CAK [of the CAN converter.
Le procède de détermination des fréquences de coupure par filtres passe-bande Ft]_ a Ft^ implemente dans le contrôleur de filtre CF est analogue au procède de détermination des gains variables, et attribue successivement chacune de J valeurs de fréquence de coupure prédéterminées aux fréquences de coupure de chacun des filtres passe-bande Fti a Ft^ afin que toutes les combinaisons de valeurs de fréquence de coupure prédéterminées soient testées par le contrôleur de gain CG. Le procède de détermination des fréquences de coupure comprend également N boucles de récurrence incluses les unes dans les autres et relatives respectivement aux variations des fréquences de coupure des filtres passe-bande Ft]_ a Ft^. Dans la dernière boucle exécutée incluse dans toutes les autres boucles, des valeurs des paires de fréquences de coupure distinctes bornant les bandes passantes des filtres Fti a FtN sont sélectionnées pour obtenir un débit total maximal Dτ = DTmax et une amplitude amp(Pχ) du signal SAS inférieure a la limite d'amplitude LIMCAN du convertisseur CAN. Le débit et l'amplitude sont détermines notamment en fonction des gains fixes prédéfinis des N étages de sélection et des puissances des porteuses filtrées dans chaque étage de sélection selon les fréquences de coupure variables testées.The method of determining the bandpass filter cut-off frequencies is implemented in the filter controller CF and is analogous to the variable gain determination process, and successively assigns each of predetermined cutoff frequency values to the frequencies. to cut each of the bandpass filters Fti to Ft ^ so that all combinations of predetermined cutoff frequency values are tested by the CG gain controller. The method for determining the cut-off frequencies also comprises N recursive loops included in each other and relating respectively to the variations of the cut-off frequencies of the band-pass filters Ft] _ a Ft ^. In the last loop executed included in all the other loops, values of the pairs of distinct cut-off frequencies bounding the passbands of the filters Fti to Ft N are selected to obtain a maximum total bit rate D τ = D Tmax and an amplitude amp (Pχ ) of the SAS signal less than the CAN CAN amplitude limit of the CAN converter. The flow rate and the amplitude are determined in particular according to the predefined fixed gains of the N stages of selection and the powers of the filtered carriers in each selection stage according to the variable cutoff frequencies tested.
Le dispositif sélectif de pondération de puissance DPP peut aussi fonctionner en combinant les deux réalisations précédemment exposées comportant pour chaque étage de sélection un filtre passe-bande ayant une bande de fréquence variable et un pondérateur de puissance appliquant un gain variable.The selective power weighting device DPP can also operate by combining the two previously described embodiments comprising for each selection stage a bandpass filter having a variable frequency band and a power weighting applying a variable gain.
Le dispositif sélectif de pondération de puissance DPP peut être intègre dans le dispositif de réception du système de communication comme présente dans les exemples précédents, ou bien avec un coupleur en entrée être distinct de celui-ci. The selective power weighting device DPP may be integrated in the receiving device of the communication system as present in the preceding examples, or with an input coupler be distinct from it.

Claims

REVENDICATIONS
1 - Procède de réception d'un signal analogique multiporteuse (SA) par un dispositif de réception (DRa) ayant un convertisseur analogique-numérique (CAN) , mettant en œuvre :1 - Method for receiving a multicarrier analog signal (SA) by a receiving device (DRa) having an analog-to-digital converter (ADC), implementing:
- N filtrages passe-bande simultanés du signal analogique respectivement en N signaux analogiques filtres ( SAFi-SAF^) incluant chacun au moins une porteuse (Fp]_-Fpχ),Simultaneous bandpass filtering of the analog signal respectively in N analog filter signals (SAFi-SAF) each including at least one carrier (Fp] _-Fpχ),
- N pondérations par N gains (GV^-GVN, GFχ-GFN) respectivement des N signaux analogiques filtres, et une sommation des signaux analogiques filtres pondères en un signal analogique somme (SAS), caractérise en ce qu'il met en outre en œuvre : une détermination de l'une des caractéristiques des filtrages et des pondérations en fonction d'un débit total maximal (Dτmax) du signal analogique somme (SAS) et de l'amplitude (amp (PiO) du signal analogique somme adaptée a une limite d'amplitude (LIM^N) du convertisseur analogique- numérique (CAN) .N weightings by N gains (GV ^ -GV N , GFχ-GF N ) respectively of the N analog filter signals, and a summation of the weighted analog filter signals into an analog signal sum (SAS), characterized in that it sets in addition: a determination of one of the characteristics of the filterings and weightings as a function of a maximum total flow rate (Dτ max ) of the sum analog signal (SAS) and of the amplitude (amp (PiO) of the analog signal sum adapted to an amplitude limit (LIM ^ N) of the analog-to-digital converter (ADC).
2 - Procède conforme a la revendication 1, selon lequel les porteuses (Fpχ-Fpχ) du signal analogique2 - Method according to claim 1, wherein the carriers (Fpχ-Fpχ) of the analog signal
(SA) ont des fréquences sélectionnées par les N filtrages passe-bande simultanés en fonction respectivement de N bandes de fréquence fixes afin de produire les N signaux filtres (SAFχ-SAFN) , et les N gains (GVχ-GVN) appliques respectivement aux signaux filtres sont variables et détermines en fonction du débit total maximal (DTmax) du signal analogique somme (SAS) et de l'amplitude (amp(Pτ)) du signal analogique somme adaptée a la limite d'amplitude (LIMCAN) . 3 - Procède conforme a la revendication 1, selon lequel les porteuses (Fpχ-Fpj) du signal analogique (SA) ont des fréquences sélectionnées selon N filtrages passe-bande simultanés en fonction respectivement de N bandes de fréquence variables déterminées en fonction de gains respectifs fixes (GF]_-GFN) a appliquer sur les signaux filtres (SAFi- SAFN), du débit total maximal (Oτmaχ) du signal analogique somme (SAS) et de l'amplitude (amp(Pτ)) du signal analogique somme adaptée a la limite d'amplitude (LIMςAN) .(SA) have frequencies selected by the N simultaneous bandpass filtering as a function respectively of N fixed frequency bands to produce the N filter signals (SAFχ-SAF N ), and the N gains (GVχ-GV N ) applied respectively to the filter signals are variable and determined according to the maximum total flow (D Tmax ) of the sum analog signal (SAS) and the amplitude (amp (Pτ)) of the analog signal sum adapted to the amplitude limit ( CAN LIM) . 3 - Method according to claim 1, wherein the carriers (Fpχ-Fpj) of the analog signal (SA) have frequencies selected according to N simultaneous bandpass filtering respectively as a function of N variable frequency bands determined according to respective gains fixed (GF] _- GFN) to apply on the filter signals (SAF- SAF N ), the maximum total flow rate (Oτmaχ) of the analog signal sum (SAS) and the amplitude (amp (P τ )) of the analog signal sum adapted to the amplitude limit (LIMς AN ).
4 - Dispositif de réception (DRa) d'un signal analogique multiporteuse (SA) comprenant un convertisseur analogique-numérique, N filtres passe-bande (Ftχ-FtN) pour filtrer le signal analogique respectivement en N signaux analogiques filtres (SAFI-SAFN) incluant chacun au moins une porteuse (Fp1-Fp1),4 - Reception device (DRa) of a multicarrier analog signal (SA) comprising an analog-to-digital converter, N bandpass filters (Ftχ-Ft N ) for filtering the analog signal respectively in N analog filter signals (SAFI-SAFN ) each including at least one carrier (Fp 1 -Fp 1 ),
N pondérateurs de puissance (Pdi-Pd^) pour pondérer par N gains (GVχ-GVtj, GFχ-GFN) respectivement les N signaux analogiques filtres, et un sommateur (SM) pour sommer les signaux analogiques filtres pondères en un signal analogique somme (SAS) , caractérise en ce qu'il comprend en outre un moyen (CG ; CF) pour déterminer au moins l'une des caractéristiques des filtres et des pondérateurs en fonction d'un débit total maximal (D<rmax) ^u signal analogique somme (SAS) et de l'amplitude (amp(Pχ)) du signal analogique somme propre a être adaptée a une limite d'amplitude (LIMCAN) du convertisseur analogique-numérique (CAN) . 5 - Dispositif conforme à la revendication 4, dans lequel les N filtres passe-bande (Fti-Ft^) sont aptes à filtrer les porteuses (Fpj-Fpj) du signal analogique (SA) en fonction respectivement de N bandes de fréquence fixes afin de produire les N signaux filtrés ( SAF]_-SAFN) , les N pondérateurs de puissance (Pdχ-PdN) ont respectivement N gains variables (GVJ-GVN) à appliquer sur les signaux filtrés, et ledit moyen (CG) est apte à déterminer les N gains variables en fonction du débit total maximal (DTmax) du signal analogique sommé (SAS) et de l'amplitude (amp(Pτ)) du signal analogique sommé adaptée à la limite d'amplitude (LIMQAN) •N power weighting (Pdi-Pd ^) for weighting by N gains (GVχ-GV t j, GFχ-GF N ) respectively the N analog filter signals, and a summator (SM) for summing the weighted analog signal signals into a signal analogue sum (SAS), characterized in that it further comprises a means (CG; CF) for determining at least one of the characteristics of the filters and weightings according to a maximum total flow rate (D <r max ) The analog signal sum (SAS) and the amplitude (amp (Pχ)) of the analog sum signal have to be adapted to an amplitude limit (LIMCAN) of the analog-to-digital converter (ADC). 5 - Device according to claim 4, wherein the N bandpass filters (Fti-Ft ^) are able to filter the carriers (Fpj-Fpj) of the analog signal (SA) as a function respectively of N fixed frequency bands so to produce the N filtered signals (SAF) _- SAFN), the N power weighting units (Pdχ-Pd N ) have respectively N variable gains (GVJ-GV N ) to be applied to the filtered signals, and said means (CG) is capable of determining the N variable gains as a function of the maximum total rate (D Tmax ) of the summed analog signal (SAS) and the amplitude (amp (Pτ)) of the summed analog signal adapted to the amplitude limit (LIMQAN) •
6 - Dispositif conforme à la revendication 4, dans lequel les N filtres passe-bande (Ftχ-Ftfj) sont aptes à filtrer les porteuses (Fpχ-Fpi) du signal analogique (SA) en fonction respectivement de N bandes de fréquence variables, et ledit moyen (CF) est apte à déterminer les N bandes de fréquence variables en fonction de gains respectifs fixes (GFj- GF^) à appliquer respectivement sur les signaux filtrés (SAFI-SAFN) par les N pondérateurs de puissance (Pd]_-Pdjj), du débit total maximal (DTmax) du signal sommé (SAS) et de l'amplitude (amp(Pτ)) du signal analogique sommé adaptée à la limite d'amplitude (LIM^AN) • 6 - Device according to claim 4, wherein the N bandpass filters (Ftχ-Ft f j) are able to filter the carriers (Fpχ-Fpi) of the analog signal (SA) as a function respectively of N variable frequency bands , and said means (CF) is able to determine the N variable frequency bands as a function of respective fixed gains (GF 1 -GF 1) to be applied respectively to the filtered signals (SAFI-SAFN) by the N power weighting (Pd) _-Pdjj), the maximum total rate (D Tmax ) of the summed signal (SAS) and the amplitude (amp (P τ )) of the summed analog signal adapted to the amplitude limit (LIM ^ AN) •
PCT/FR2009/050155 2008-02-04 2009-02-02 Power weighting of a multicarrier signal on reception in a communication system WO2009101315A1 (en)

Applications Claiming Priority (2)

Application Number Priority Date Filing Date Title
FR0850675 2008-02-04
FR0850675A FR2927202A1 (en) 2008-02-04 2008-02-04 POWER PONDERATION OF A MULTI-CARRIER SIGNAL TO RECEPTION IN A COMMUNICATION SYSTEM.

Publications (1)

Publication Number Publication Date
WO2009101315A1 true WO2009101315A1 (en) 2009-08-20

Family

ID=40011143

Family Applications (1)

Application Number Title Priority Date Filing Date
PCT/FR2009/050155 WO2009101315A1 (en) 2008-02-04 2009-02-02 Power weighting of a multicarrier signal on reception in a communication system

Country Status (2)

Country Link
FR (1) FR2927202A1 (en)
WO (1) WO2009101315A1 (en)

Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2011156794A3 (en) * 2010-06-11 2013-02-28 The Regents Of The University Of California Synthetic pathways for biofuel synthesis
WO2015028582A2 (en) 2013-08-29 2015-03-05 Dsm Ip Assets B.V. Glycerol and acetic acid converting yeast cells with improved acetic acid conversion

Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP0093651A1 (en) * 1982-05-04 1983-11-09 Thomson-Csf Multiple-carrier receiver protected against jamming
EP0704986A2 (en) * 1994-09-30 1996-04-03 AT&T Corp. CDMA cellular communications with multicarrier signal processing
EP0762661A2 (en) * 1995-08-07 1997-03-12 AT&T IPM Corp. Channelized multi-carrier signal processor
US20070164823A1 (en) * 2006-01-19 2007-07-19 Mediatek Inc. Automatic gain control apparatus

Patent Citations (4)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP0093651A1 (en) * 1982-05-04 1983-11-09 Thomson-Csf Multiple-carrier receiver protected against jamming
EP0704986A2 (en) * 1994-09-30 1996-04-03 AT&T Corp. CDMA cellular communications with multicarrier signal processing
EP0762661A2 (en) * 1995-08-07 1997-03-12 AT&T IPM Corp. Channelized multi-carrier signal processor
US20070164823A1 (en) * 2006-01-19 2007-07-19 Mediatek Inc. Automatic gain control apparatus

Cited By (2)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
WO2011156794A3 (en) * 2010-06-11 2013-02-28 The Regents Of The University Of California Synthetic pathways for biofuel synthesis
WO2015028582A2 (en) 2013-08-29 2015-03-05 Dsm Ip Assets B.V. Glycerol and acetic acid converting yeast cells with improved acetic acid conversion

Also Published As

Publication number Publication date
FR2927202A1 (en) 2009-08-07

Similar Documents

Publication Publication Date Title
EP0950306B1 (en) Method and device for formatting a clipping noise in a multicarrier modulation
US8861661B2 (en) Interference wave suppressing apparatus, relay apparatus, relay system, and interference wave suppressing method
EP1583306B1 (en) Multicarrier receiver with estimation of channel time variance
EP0333167A1 (en) Receiver for a space diversity radio transmission system comprising a diversity combiner
WO2009101315A1 (en) Power weighting of a multicarrier signal on reception in a communication system
EP0814575A1 (en) Equalizer-combiner for a diversity receiver, receiver and corresponding diversity reception method
EP3281375B1 (en) Method of reducing the peak factor of a multichannel emission by intelligent and adaptive clipping/filtering
EP1311062B1 (en) Method for optimising the efficiency of an amplifier for multicarrier signals
EP1311097A1 (en) Method for clipping a composite multicarrier modulated signal
EP2668719B1 (en) Weighting filter, and corresponding methods and devices for transmitting and receiving an analog signal and computer program
CA2060413C (en) Interfering signal detection method and device for a digital data demodulator
FR2927203A1 (en) Analog-digital conversion method for large band multicarrier analog signal in e.g. power line carrier communication system, involves weighting filtered analog signals by applying gains, and adding digital signals into unique signal
FR2826208A1 (en) System and method for transmitting an audio or phonic signal, by utilizing a digital modem for transmitting analogue signals
EP3125486B1 (en) Papr reduction of a single-carrier transmission signal
WO2003001689A2 (en) Wireless signal amplification system and a television signal decoder comprising one such system
EP0317862B1 (en) Device for enhancing hearing comfort by suppressing the transient phenomena in a reception chain of a narrow band fm/pm equipment, particularly in radiotelephony
EP2380322A1 (en) Method and device for reducing quantification noise for transmitting a multi-carrier signal
EP1271473B1 (en) System and method for PRE-AND POST-PROCESSING of an audio signal for transmission over a strongly distorted channel
WO2012066209A1 (en) Data reception device, and corresponding reception method, computer program, and recording medium
EP0333042A1 (en) Receiver for a space diversity radio transmission system
EP0003203B1 (en) Method and arrangement for measuring distortions caused by the transmission medium in wideband digital radio links
FR3090247A1 (en) Method and device for the reduction of the multichannel crest factor
WO2006114513A1 (en) Method for transmitting a signal modulated with high amplitude dynamics, corresponding transmitter and receiver
EP1708374A1 (en) Single frequence repeater for terrestrial digital television networks
EP0316839A1 (en) Listening comfort enhancer in the presence of carrier fading in an FM/PM narrow-band receiver

Legal Events

Date Code Title Description
NENP Non-entry into the national phase

Ref country code: DE